1. Field of the Invention
The present invention relates to amplifiers for use as part of a closed loop system such as low dropout voltage regulators, and in particular, to amplifiers for use in such systems using Miller-effect feedback.
2. Description of the Related Art
Closed loop systems, by definition, use various forms of feedback, generally for purposes of stabilization of circuit operation. One example of such a closed loop system is a low dropout voltage regulator (LDO). (The following discussion will be within the context of an LDO, but it should be understood that the principles and advantages of the present invention can be realized and implemented in other forms of closed loop systems as well.) As is well known in the art, an LDO is a closed loop system that provides current to a load at a specific voltage, with such load voltage typically being very close in value to the overall system power supply voltage. Typically such an LDO is a self-contained system that is placed onto a printed circuit board as part of a larger host system, with the LDO generally being a mixture of on-chip and off-chip components.
Referring to FIG. 1, for example, the components providing an LDO function will frequently include an integrated circuit, as noted above, mounted on a printed circuit board within a host system (not shown). The integrated circuit will include the on-chip error amplifier TCA (e.g., a transconductance amplifier), along with its reference voltage source and an internal load. External to the integrated circuit and also resident on the printed circuit board, will be a power transistor Q1 (typically a PNP bipolar junction transistor) and an external filter capacitor. The power transistor Q1 serves as the regulating circuit element between the main power supply VDD and the regulated power supply voltage source Vreg. The filter capacitor external filters the regulated voltage Vreg, shunting any spurious or noise components to the negative power supply rail VSS (or ground GND). The error amplifier TCA must be able to drive the base terminal of the power transistor Q1 such that the transistor Q1 is fully on. Accordingly, this puts some restrictions on the design of the output stage of the error amplifier TCA.
Referring to FIGS. 2A and 2B, a typical output stage for the error amplifier TCA is a P-type metal oxide semiconductor field effect transistor (PMOSFET) M11. The gate terminal of transistor M11 is driven by an input voltage Vin and, in turn, the source terminal of transistor M11 provides the base voltage VB for the base terminal of the power transistor Q1. Further in turn, the power transistor Q1 provides the current for the load Rload across which the regulated output voltage Vreg appears and is filtered by the external filter capacitance which includes a capacitive component Cload and an effective series resistance Resr (discussed in more detail below).
This circuit arrangement is chosen to provide good control of the DC gain from input voltage Vin to intermediate voltage VB, as well as good control over the transfer function pole associated with the base terminal of transistor Q1. As indicated in the circuit model of FIG. 2B, there are two poles and two zeros in the transfer function for this system. The first pole P1 and zero Z1 are associated with the output node where the output voltage Vreg appears, while the second pole P2 and zero Z2 are associated with the base terminal of transistor Q1.
Pole 1 P1 is a function of the load capacitance Cload and load resistance Rload. With the load capacitance Cload fixed, this pole P1 becomes a linear function of the load resistance Rload. Pole 2 is a function of the input capacitance Cpi and resistance Rpi (as components of the input impedance of the transistor Q1 (in parallel with the output impedance gm*Vgs of source follower transistor M11). In accordance with well known transistor principles, this transistor input resistance Rpi and capacitance Cpi are, at least to a first order approximation, linearly dependent on the collector current IC of transistor Q1, while the transconductance gm of the PMOS transistor M11 is square law dependent on the collector current IC (due to its relationship to its base current IB of transistor Q1, which is equal to the drain current ID of transistor M11). If transistor M11 is scaled such that the input resistance Rpi of transistor Q1 is much lower than the inverse of the transconductance of transistor M11, then pole P2 will stay relatively constant over a broad range of load resistance Rload. Further, as a practical matter, the size of transistor M11 often cannot be so large that it stays in saturation over the entire range of load resistance Rload. Therefore, at load currents below the saturation level, transistor M11 will behave resistively, thereby keeping the pole P2 relatively stable.
The voltage gain from the input Vin at the gate terminal of transistor M11 to the base voltage VB at the base terminal of transistor Q1 will increase as a function of the transistor Q1 collector current IC raised to a power of 1.5 when the load current is reduced until transistor M11 transitions into its linear region of operation, where it will then increase linearly to a maximum of unity. Since the output voltage Vreg will be fixed, the output or load current necessarily has a linear relationship with the load resistance Rload. Accordingly, the voltage gain from the base voltage Vb to the collector, or output, voltage Vreg should remain constant. As a practical matter, however, the emitter resistance of transistor Q1, albeit small, will cause the transconductance of transistor Q1 to degenerate for large collector current IC.
The second zero Z2 is associated with the feed forward path provided by parasitic capacitance CMU of transistor Q1. For low load impedances, this zero Z2 goes to a very high frequency. The first zero Z1 is associated with the effective series resistance (RESR) of the load capacitor Cload. This resistance will remain substantially constant. The load resistance Rload is normally assumed to be nominally resistive. With a large load capacitance Cload and low load resistance Rload, the output node produces a high frequency pole. However, under high load resistance Rload, this output node pole becomes significantly lower. Accordingly, the circuit cannot be compensated at the output node because a low frequency dominant pole does not always exist. Similarly, to cover the entire range of possible load resistance values, a very large capacitance Cload would be required to compensate the high impedance node within the error amplifier. Such a large capacitance would require a very large current to slew. However, to produce a high DC gain, the output current must remain low. These two requirements conflict with each other, plus the required capacitor would be too large for a practical design.
A more practical solution has been to use Miller feedback, i.e., a feedback capacitance Cm between the load terminal and internal terminal of the error amplifier. However, it has been shown that traditional Miller feedback can severely degrade the power supply rejection ratio (PSRR) of the circuit. On the other hand, it has also been shown that connecting the Miller feedback capacitance back to a low impedance node of the amplifier rather than connecting it to a high impedance node can provide the same Miller capacitance gain while significantly improving the PSRR.
Referring to FIG. 3, one example of an LDO uses a complementary MOSFET (CMOSFET) folded cascode amplifier stage (P-MOSFETs M1, M2, M7, M8, M9, M10 and M12, and N-MOSFETs M3, M4, M5 and M6) to provide high DC gain, a wide output voltage swing close to the positive power supply rail VDD, and to allow a low input reference voltage Vref of 1.2 volts (e.g., provided by a bandgap voltage source). This cascode stage drives the output transistor M11 of the error amplifier, which in turn, drives the external power transistor Q1, as discussed above. The Miller feedback capacitance couples the output terminal at the collector of the power transistor Q1 to a low impedance node of the error amplifier at the source terminal of cascode transistor M6. Transistors M3 and M4 are biased by a fixed bias voltage Vbn1, transistors M5 and M6 are biased by another fixed bias voltage Vbn2, and transistors M9, M10 and M12 are biased by still another fixed bias voltage Vbp1. The gate of transistor M1 receives the sense voltage which is tapped off of a resistive voltage divider Rtop, Rbottom at the output, while the gate terminal of transistor M2 receives the reference voltage from the reference voltage source (FIG. 1).
Referring to FIG. 4, this system includes two closed loops: the Miller loop and the DC loop. The Miller loop begins at the top of the load resistor Rload, follows through the Miller feedback capacitance CM, through the channel (from source to drain terminals) of transistor M6 into the gate and out the source terminals of transistor M11, and into the base and out the collector terminals of transistor Q1 back to the top of the load resistor Rload. The DC loop begins at the sense voltage terminal and passes through transistors M1 and M2 to the source terminal of transistor M6 and into the Miller loop which ultimately conveys the signal back to the sense terminal.
Referring to FIG. 5, the DC loop terminates into the Miller loop at the source terminal at transistor M6, as noted above, and can be modeled as an ideal transconductor as depicted. This transconductor gm*(Vregxe2x88x92Vref) pushes current into the Miller loop at the source terminal of transistor M6. The resultant voltage appearing at the collector terminal of transistor Q1 is divided down by the voltage divider Rtop, Rbottom to generate the feedback sense voltage Vsense. Since the transconductor output current develops a voltage across impedance in the Miller loop, the magnitude and phase of that impedance will determine the gain and phase of the DC loop.
Referring to FIG. 6, in a somewhat simplified view of the Miller loop, five transfer function poles and four transfer function zeros can be identified. Poles 1 and 2 and zeros 1 and 2 are as discussed above in connection with FIG. 2B. Zero 3 need not be considered for the purposes of this discussion, while zero 4 is at DC. Pole 5 is at a very high frequency because of the low impedance associated with the source terminal of transistor M10. Pole 3 is at a low frequency and is a function of a ratio of the Miller feedback capacitance CM, the parasitic capacitance associated with the junction of the gate terminal of transistor M11 and drain terminals of transistors M6 and M10, and the high impedance of such junction as reflected back to the source terminal of transistor M6. Pole 4 is at a frequency defined by a ratio of the transconductants of transistor M6 to the capacitance associated with the drain terminal of transistor M6.
In accordance with the presently claimed invention, circuitry includes Miller-effect feedback for use as part of a closed loop system such as a low dropout voltage regulator that provides current to a load at a specified voltage close in value to the power supply voltage. Various aspects of the presently claimed invention include using, within the Miller-effect feedback loop: a buffer amplifier to reduce loading effects upon an internal high impedance circuit node, output compensation circuitry to introduce a transfer function pole for substantially canceling a transfer function zero associated with external load circuitry; and Miller-effect compensation circuitry to introduce a transfer function zero for substantially canceling a transfer function pole associated with the Miller-effect feedback.
In accordance with one embodiment of the presently claimed invention, an integrated circuit with a buffer amplifier inside a Miller-effect feedback loop for reducing loading on an internal high impedance circuit terminal includes first amplification circuitry, buffer amplification circuitry and second amplification circuitry. The first amplification circuitry includes an internal terminal and a first amplifier output terminal, wherein: the internal terminal has an internal terminal impedance associated therewith; the first amplifier output terminal has a first amplifier output terminal impedance associated therewith; the internal terminal impedance is substantially lower than the first amplifier output terminal impedance; and the first amplification circuitry is adapted to couple via the internal terminal to a Miller-effect feedback capacitance. The buffer amplification circuitry includes buffer input and output terminals, wherein the buffer input terminal is coupled to the first amplifier output terminal. The second amplification circuitry includes a second amplifier input terminal coupled to the buffer output terminal, wherein: the second amplifier input terminal has a second amplifier input terminal impedance associated therewith; the second amplifier input terminal impedance is substantially lower than the first amplifier output terminal impedance; and the second amplification circuitry is adapted to couple to further amplification circuitry that includes the Miller-effect feedback capacitance.
In accordance with another embodiment of the presently claimed invention, an integrated circuit with a buffer amplifier inside a Miller-effect feedback loop for reducing loading on an internal high impedance circuit node includes first amplifier means, buffer amplifier means and second amplifier means. The first amplifier means is for receiving via an internal node a feedback signal via a Miller-effect feedback capacitance and providing via a first amplifier output node a first amplified signal, wherein: the internal node has an internal node impedance associated therewith; the first amplifier output node has a first amplifier output node impedance associated therewith; and the internal node impedance is substantially lower than the first amplifier output node impedance. The buffer amplifier means is for receiving and buffering the first amplified signal and providing a buffered signal. The second amplifier means is for receiving via a second amplifier input node and amplifying the buffered signal and providing a second amplified signal for further amplifier means including the Miller-effect feedback capacitance, wherein: the second amplifier input node has a second amplifier input node impedance associated therewith; and the second amplifier input node impedance is substantially lower than the first amplifier output node impedance.
In accordance with still another embodiment of the presently claimed invention, an integrated circuit with a compensation circuit inside a Miller-effect feedback loop includes first amplification circuitry, second amplification circuitry and compensation circuitry. The first amplification circuitry includes an internal terminal and a first amplifier output terminal, wherein: the first amplifier output terminal has a first amplifier output terminal impedance associated therewith; and the first amplification circuitry is adapted to couple via the internal terminal to a Miller-effect feedback capacitance. The second amplification circuitry includes a second amplifier input terminal having a second amplifier input terminal impedance associated therewith, wherein: the second amplifier input terminal impedance is substantially lower than the first amplifier output terminal impedance; and the second amplification circuitry is adapted to couple to further amplification circuitry that includes the Miller-effect feedback capacitance and a shunt capacitance having an effective series resistance associated therewith. The compensation circuitry is coupled between the first amplification circuitry and the second amplification circuitry. The first amplification circuitry, the compensation circuitry and the second amplification circuitry together, when coupled to the further amplification circuitry, have a circuit transfer function associated therewith that includes: a transfer function zero associated with at least a portion of the further amplification circuitry; and a transfer function pole associated with the compensation circuitry. The transfer function zero and pole are at substantially equal frequencies.
In accordance with still another embodiment of the presently claimed invention, an integrated circuit with a compensation circuit inside a Miller-effect feedback loop includes first amplifier means, compensation means and second amplifier means. The first amplifier means is for receiving via an internal node a feedback signal via a Miller-effect feedback capacitance and providing via a first amplifier output node a first amplified signal, wherein the first amplifier output node has a first amplifier output node impedance associated therewith. The compensation means is for compensating the first amplified signal and providing a compensated signal. The second amplifier means is for receiving via a second amplifier input node and amplifying the compensated signal and providing a second amplified signal for further amplifier means including the Miller-effect feedback capacitance and a shunt capacitance having an effective series resistance associated therewith, wherein: the second amplifier input node has a second amplifier input node impedance associated therewith; and the second amplifier input node impedance is substantially lower than the first amplifier output node impedance. The first amplifier means, the compensation means and the second amplifier means together, when coupled to the further amplifier means, have a transfer function associated therewith that includes: a transfer function zero associated with at least a portion of the further amplifier means; and a transfer function pole associated with the compensation means. The transfer function zero and pole are at substantially equal frequencies.
In accordance with yet another embodiment of the presently claimed invention, an integrated circuit with a plurality of compensation circuits inside a Miller-effect feedback loop includes input amplification circuitry, buffer amplification circuitry, Miller-effect compensation circuitry, output amplification circuitry and output compensation circuitry. The input amplification circuitry includes an internal terminal and an input amplifier output terminal, wherein: the input amplifier output terminal has an input amplifier output terminal impedance associated therewith; and the input amplification circuitry is adapted to couple via the internal terminal to a Miller-effect feedback capacitance. The buffer amplification circuitry includes a buffer input terminal coupled to the input amplifier output terminal. The Miller-effect compensation circuitry is coupled between the input amplification circuitry and the buffer amplification circuitry. The output amplification circuitry includes an output amplifier input terminal and is adapted to couple to further amplification circuitry that includes the Miller-effect feedback capacitance and a shunt capacitance having an effective series resistance associated therewith. The output compensation circuitry is coupled between the buffer and output amplification circuitry. The input amplification circuitry, the Miller-effect compensation circuitry, the buffer amplification circuitry, the output compensation circuitry, and the output amplification circuitry together, when coupled to the further amplification circuitry, have a circuit transfer function associated therewith that includes: a first transfer function zero associated with the Miller-effect compensation circuitry; a second transfer function zero associated with at least a portion of the further amplification circuitry; a first transfer function pole associated with the Miller-effect feedback capacitance; and a second transfer function pole associated with the output compensation circuitry. The first transfer function zero and pole are at first substantially equal frequencies, and the second transfer function zero and pole are at second substantially equal frequencies.
In accordance with yet another embodiment of the presently claimed invention, an integrated circuit with a plurality of compensation circuits inside a Miller-effect feedback loop includes input amplifier means, Miller-effect compensation means, buffer amplifier means, output compensation means and output amplifier means. The input amplifier means is for receiving via an internal node a feedback signal via a Miller-effect feedback capacitance and providing via an input amplifier output node a first amplified signal, wherein the input amplifier output node has an input amplifier output node impedance associated therewith. The Miller-effect compensation means is for compensating the first amplified signal and providing a first compensated signal. The buffer amplifier means is for receiving and buffering the first compensated signal and providing a buffered signal. The output compensation means is for compensating the buffered signal and providing a second compensated signal. The output amplifier means is for receiving via an output amplifier input node and amplifying the second compensated signal and providing a second amplified signal for further amplifier means including the Miller-effect feedback capacitance and a shunt capacitance having an effective series resistance associated therewith. The input amplifier means, the Miller-effect compensation means, the buffer amplifier means, the output compensation means and the output amplifier means together, when coupled to the further amplifier means, have a transfer function associated therewith that includes: a first transfer function zero associated with the Miller-effect compensation means; a second transfer function zero associated with at least a portion of the further amplifier means; a first transfer function pole associated with the Miller-effect feedback capacitance; and a second transfer function pole associated with the output compensation means. The first transfer function zero and pole are at first substantially equal frequencies, and the second transfer function zero and pole are at second substantially equal frequencies.